MOS capacitors structures for variable capacitor arrays and methods of forming the same

ABSTRACT

A capacitor structure is described. A capacitor structure including a substrate; a source/drain region formed in the substrate to form an active area having an active area width; and a plurality of gates formed above the substrate. The source/drain region having a reflection symmetry. Each of the plurality of gates having a gate width. The gate width is configured to be less than said active area width. And, the plurality of gates are formed to have reflection symmetry.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application No.62/140,319 filed on Mar. 30, 2015, which is hereby incorporated byreference in its entirety.

FIELD

Embodiments of the invention relate to electronic systems and, inparticular, to variable capacitors.

BACKGROUND

A wireless device such as a smart phone, tablet, or laptop computer cancommunicate over multiple frequency bands using one or more common orshared antennas. A desire to transmit at wider bandwidth and/or overdifferent communications networks has increased a demand for the numberof bands that a wireless device can communicate over. For example, awireless device may be specified to operate using one or more of avariety of communications standards including, for example, GSM/EDGE,IMT-2000 (3G), 4G, Long Term Evolution (LTE), Advanced LTE, IEEE 802.11(Wi-Fi), Mobile WiMAX, Near Field Communication (NFC), GlobalPositioning System (GPS), GLONASS, Galileo, Bluetooth, and the like.Proprietary standards can also be applicable. The complexities ofmulti-band communication can be further exacerbated in configurations inwhich the wireless device is specified to use carrier aggregation.

SUMMARY

A capacitor structure is described. A capacitor structure including asubstrate; a source/drain region formed in the substrate to form anactive area having an active area width; and a plurality of gates formedabove the substrate. The source/drain region having a reflectionsymmetry. Each of the plurality of gates having a gate width. The gatewidth is configured to be less than said active area width. And, theplurality of gates are formed to have reflection symmetry.

Other features and advantages of embodiments of the present inventionwill be apparent from the accompanying drawings and from the detaileddescription that follows.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention are illustrated by way of exampleand not limitation in the figures of the accompanying drawings, in whichlike references indicate similar elements and in which:

FIG. 1 is a schematic plan view of one embodiment of a metal oxidesemiconductor (MOS) capacitor structure for a variable capacitor arrayaccording to an embodiment;

FIG. 2A is a schematic diagram of one embodiment of a radio frequency(RF) system;

FIG. 2B is a schematic diagram of another embodiment of an RF system;

FIG. 2C is a schematic diagram of another embodiment of an RF system;

FIG. 3 illustrates a schematic diagram of a programmable filteraccording to one embodiment;

FIG. 4A illustrates a schematic diagram of one embodiment of an RFsignal processing circuit according to an embodiment;

FIG. 4B illustrates a schematic diagram of another embodiment of an RFsignal processing circuit according to an embodiment;

FIG. 5 illustrates a schematic diagram of an IC according to anotherembodiment;

FIGS. 6A and 6B illustrate graphs of two examples of capacitance versusbias voltage;

FIG. 7 illustrates a schematic diagram of an IC according to anotherembodiment;

FIG. 8 illustrates a schematic diagram of an IC according to anotherembodiment;

FIG. 9A illustrates a circuit diagram of a variable capacitor cellaccording to one embodiment;

FIG. 9B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 10A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 10B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 11A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 11B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 12A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 12B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 13A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 13B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 14A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 14B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 15A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 15B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 16A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 16B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 17A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 17B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 18A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 18B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 19A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 19B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 20A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 20B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 21A illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 21B illustrates a circuit diagram of a variable capacitor cellaccording to another embodiment;

FIG. 22 is a schematic diagram of a cross section of an IC according toone embodiment;

FIG. 23A is a cross section of a MOS capacitor according to oneembodiment; and

FIG. 23B is a cross section of a MOS capacitor according to anotherembodiment.

DETAILED DESCRIPTION

Disclosed herein are MOS capacitor structures for variable capacitorarrays.

In certain embodiments, the MOS capacitor structures can be fabricatedusing silicon on insulator (SOI) processes. Thus, the MOS capacitorstructures can be included in an integrated circuit (IC) that is in anSOI substrate. For example, the integrated circuit can include a supportsubstrate, an insulator layer (for example, a buried oxide layer) overthe support substrate, and a device layer over the insulator layer. TheMOS capacitors can include source and drain diffusion regions that areformed in the device layer.

MOS capacitors can be included in variable capacitor array that includesa plurality of MOS variable capacitor cells. In certain configurations,the plurality of MOS variable capacitor cells can include pairs of MOScapacitors that are implemented in anti-series and/or anti-parallelconfigurations. Examples of MOS variable capacitor arrays can be asdescribed in Ser. No. 14/559,783 and in U.S. Patent Publication No.2014/0354348, now U.S. Pat. No. 9,086,709, each of which are herebyexpressly incorporated by reference herein in their entirety.

FIG. 1 is a schematic plan view of one embodiment of a metal oxidesemiconductor (MOS) capacitor structure for a variable capacitor array.

The illustrated MOS capacitor structure 1000 includes gates 1040 that donot extend across the full source/drain diffusion region 1020. Such adevice can be referred to herein as a flow device. Configuring the MOScapacitor structure 1000 in this manner can provide a resistive DC paththrough source/drain diffusion which can be used to bias the device.

The one or more source/drain regions 1020 are formed in a substrate,such as an SOI substrate, using doping techniques including those knownin the art. The plurality of gates 1040 are formed above the substrate,using techniques including those known in the art, such that a fingerspacing 1090 is maintained between each gate 1040 of the plurality ofgates 1040. The plurality of gates 1040 are formed above thesource/drain region 1020 to form a channel 1100 within the active areawidth 1050. The plurality of gates 1040 are interconnected using a metalinterconnect 1080 and one or more contacts 1060.

Further, for various embodiments, a plurality of gates 1040 are formedabove the substrate. For example, the plurality of gates 1040 a areformed of polysilicon formed above the substrate on an oxide layer usingtechniques including those known in the art. Such a plurality of gates1040 are formed having a gate/finger width 1030 such that the gate width1030 is configured to be less than the active area width 1050. Further,the source/drain (S/D) regions 1020 may be interconnected using a metalinterconnect. The metal interconnects are formed using techniquesincluding those known in the art. Thus, the MOS capacitor structure 1000includes a separation or gap 1070 between an end of the gate and an edgeof the source/drain diffusion region 1020.

The MOS capacitor structure 1000 can be biased using a relatively smallnumber of contacts. In certain configurations, the S/D region 1020 iscontacted using contacts 1060. The contacts 1060 are formed usingtechniques including those known in the art.

Certain MOS capacitor devices can be biased using a set of contacts foreach FET channel area (E). Including a set of contacts for each FETchannel area can increase finger-to-finger spacing (I) and an overallactive area height (D) for a given (width (W)*length (L)*number offingers) FET channel area.

In contrast, the illustrated MOS capacitor structure 1000 includes gates1040 that do not extend the full source/drain diffusion width (activearea width) 1050. This advantageously decreases active area height (D)1093 decreases the length of the input/output routes (H) 1090, therebydecreasing the series resistance of these routes and increasingQ-factor.

The MOS variable capacitor arrays implemented in accordance with theteachings herein can exhibit high densities.

Substrate coupling due to routing over the substrate can degradeQ-factor. Such substrate coupling can be proportional to area of thecircuit element and/or distance of the circuit element from thesubstrate.

In certain implementations, area reduction is in favor of active areaover polysilicon area. Active area can have a poorer substrateperformance, relative to polysilicon area.

In one embodiment, the gate and/or source/drain regions of the variablecapacitor structure exhibits reflectional symmetry. For variousembodiments, a source/drain region 1020 is formed to have a reflectionalsymmetry along one or more lines of symmetry. Further, the plurality ofgates 1040 formed above the source/drain region 1020 are formed to havea reflectional symmetry along one or more lines of symmetry.

The MOS capacitor structure of FIG. 1 can represent a portion of alarger variable capacitor array.

Additionally, the MOS capacitor structure can be adapted to include moreor fewer gate and active regions and/or different configurations ofmetallization and contacts to aid in implementing a variable capacitorarray with a desired overall performance characteristic. For instance,the MOS capacitor structure can be scaled, replicated, and/or mirroredto implement a variable capacitor array including a desired number ofand/or configuration of MOS variable capacitor cells.

FIG. 2A is a schematic diagram of one embodiment of a radio frequency(RF) system 10. The RF system 10 includes a programmable duplexer 1, anantenna 2, a receive terminal RX, and a transmit terminal TX. The RFsystem 10 can represent a portion of a wireless device, such as a smartphone. Accordingly, although not illustrated in FIG. 2A for clarity, theRF system 10 can include additional components and/or circuitry.

As shown in FIG. 2A, the programmable duplexer 1 includes a firstprogrammable filter 3 and a second programmable filter 4. The firstprogrammable filter 3 includes an input electrically connected to theantenna 2 and an output electrically connected to the receive terminalRX. The first programmable filter 3 further includes a first variablecapacitor array 5, which can be used to control a filteringcharacteristic of the first programmable filter 3, such as the locationin frequency of a passband. The second programmable filter 4 includes aninput electrically connected to the transmit terminal TX and an outputelectrically connected to the antenna 2. The second programmable filter4 further includes a second variable capacitor array 6, which can beused to control a filtering characteristic of the second programmablefilter 4.

A wireless device such as a smart phone, tablet, or laptop computer cancommunicate over multiple frequency bands using one or more common orshared antennas. A desire to transmit at wider bandwidth and/or overdifferent communications networks has increased a demand for the numberof bands that a wireless device can communicate over. For example, awireless device may be specified to operate using one or more of avariety of communications standards including, for example, GSM/EDGE,IMT-2000 (3G), 4G, Long Term Evolution (LTE), Advanced LTE, IEEE 802.11(Wi-Fi), Mobile WiMAX, Near Field Communication (NFC), GlobalPositioning System (GPS), GLONASS, Galileo, Bluetooth, and the like.Proprietary standards can also be applicable. The complexities ofmulti-band communication can be further exacerbated in configurations inwhich the wireless device is specified to use carrier aggregation.

Certain conventional wireless devices can include a multi-throw switchand a duplexer associated with each of the frequency bands, and themulti-throw switch can be used to selectively couple an antenna to aduplexer associated with a particular band. The duplexers can provideband filtering using, for example, passive filtering structures, such asa surface acoustic wave (SAW) filters and/or thin film bulk acousticresonators (FBARs). The multi-throw switch can be used to electricallycouple the antenna to a duplexer associated with a frequency band thatthe wireless device is transmitting and/or receiving over at aparticular time instance.

In the illustrated configuration, the programmable duplexer 1 can beconfigured to filter a particular frequency band by programming thefirst and second programmable filters 3, 4 using a control signal CNTL.For example, in certain embodiments, the capacitance value of the firstvariable capacitor array 5 can be controlled using the control signalCNTL to control a frequency location of a passband of the firstprogrammable filter 3, and the capacitance value of the second variablecapacitor array 6 can be controlled using the control signal CNTL tocontrol a frequency location of a passband of the second programmablefilter 4.

Accordingly, the programmable duplexer 1 can be used to provide the RFsystem 10 with multi-band capability, while avoiding a need for using amulti-throw switch and a duplexer for each frequency band. Including theprogrammable duplexer 1 in the RF system 10 can reduce insertion loss intransmit and/or receive paths by eliminating a need for a multi throwswitch. Furthermore, the programmable duplexer 1 can have smaller arearelative to a configuration including a multi-throw switch and multipleduplexers. Thus, a wireless device that includes the programmableduplexer 1 can have a smaller form factor and/or lower cost.

In the illustrated configuration, the capacitance values of the firstand second variable capacitor arrays 5, 6 can be controlled using thecontrol signal CNTL. In one embodiment, the control signal CNTL isreceived by the programmable duplexer 1 over an interface, such as aserial peripheral interface (SPI) or Mobile Industry Processor Interfaceradio frequency front end (MIPI RFFE) interface. Although two examplesof interfaces have been provided, other interfaces can be used. AlthoughFIG. 2A illustrates the first and second variable capacitor arrays 5, 6as receiving a common control signal CNTL, other configurations arepossible, such as implementations in which the first and second variablecapacitor arrays 5, 6 are controlled using separate control signals.

The first variable capacitor array 5 and/or the second variablecapacitor structure 6 can be implemented using one or more embodimentsof variable capacitor arrays described herein. Thus, the first andsecond variable capacitor arrays 5, 6 can include metal oxidesemiconductor (MOS) capacitors, which can offer enhanced performanceover certain other tunable capacitance structures. For instance, certainmicroelectromechanical systems (MEMS) capacitors can exhibit lowQ-factor, poor reliability, and/or limited tuning range. Additionally,other approaches such as coupled resonators can suffer from large sizeand/or cost, and thus can be unsuitable for certain applications,including smart phones.

FIG. 2B is a schematic diagram of another embodiment of an RF system2000 that includes an RF circuit 1500. The RF circuit 1500 includes atunable input matching network 2100 electrically connected to an RFinput IN and a tunable output matching network 2200 electricallyconnected to an RF output OUT. As shown in FIG. 2B, the tunable inputmatching network 2100 and the tunable output matching network 2200include first and second variable capacitor arrays 5, 6, respectively.

The first variable capacitor array 5 receives the control signal CNTL,which can be used to control the first variable capacitor array'scapacitance. The capacitance of the first variable capacitor array 5 canbe used to control, for example, an input impedance of the RF circuit1500 and/or to control a ratio of impedance transformation provided bythe tunable input matching network 2100. Additionally, the capacitanceof the second variable capacitor array 6 can be controlled by thecontrol signal CNTL, thereby controlling, for example, an outputimpedance of the RF circuit 1500 and/or a ratio of impedancetransformation provided by the tunable output matching network 2200.

Including the tunable input matching network 2100 and the tunable outputmatching network 2200 can enhance performance in a variety of ways, suchas improving performance under varying voltage standing wave ratio(VSWR). The first and second variable capacitor arrays 5, 6 can beimplemented in accordance with the teachings herein to provide high RFvoltage handling capabilities, high Q-factor, low insertion loss, and/orhigh linearity.

FIG. 2C is a schematic diagram of another embodiment of an RF system3000 that includes an antenna tuning circuit 3100 and an antenna 2. Theantenna tuning circuit 3100 is electrically connected between an RFterminal IN and the antenna 2.

As shown in FIG. 2C, the antenna tuning circuit 31 includes the variablecapacitor array 5, which can be controlled using the control signalCNTL. The capacitance of the variable capacitor array 5 can be used, forexample, to control an impedance transformation provided by the antennatuning circuit 3100 and/or a standing wave ratio on the RF terminal IN.

Although the RF systems of FIGS. 2A-2C illustrate various examples ofelectronic systems that can include one or more variable capacitorarrays, the variable capacitor arrays described herein can be used inother electronic systems. For example, variable capacitor arrays can beused in wide range of RF electronics, including, for example,programmable filters, programmable resonators, programmable antennatuners, programmable impedance matching networks, programmable phaseshifters, and/or programmable duplexers.

FIG. 3 is a schematic diagram of a programmable filter 20 according toone embodiment. The programmable filter 20 includes an input impedancetransformer 11, a splitter transformer 12, an RF signal processingcircuit 13, a combiner transformer 14, and an output impedancetransformer 15. The programmable filter 20 further includes an RF inputIN and an RF output OUT. For various embodiments, the programmablefilter 20 is configured as a tunable notch filter including thosedescribed herein.

The input impedance transformer 11 can receive an RF input signal on theRF input IN, and can generate an impedance transformed signal 21. Theinput impedance transformer 11 can provide an impedance transformationfrom input to output. For example, in one embodiment, the inputimpedance transformer 11 transforms an input impedance of about 50Ω toan output impedance of about R_(L), where R_(L) is less than 50Ω, forexample, 8Ω.

Transforming the input impedance of the programmable filter 20 in thismanner can result in the impedance transformed signal 21 having asmaller voltage level relative to a voltage level of the RF input signalreceived at the RF input IN. For example, when the programmable filter20 has an input impedance of about 50Ω, the voltage level of theimpedance transformed signal 21 can be smaller than the voltage level ofthe RF input signal by a factor of about √{square root over (50/R_(L))}.

The splitter transformer 12 can receive the impedance transformed signal21 from the input impedance transformer 11, and can generate N splitsignals, where N is an integer greater than or equal to 2. In theillustrated configuration, the splitter transformer 12 generates a firstsplit signal 22 a, a second split signal 22 b, and a third split signal22 c. Although an example with N=3 has been illustrated, the principlesand advantages disclosed herein are applicable to a broad range ofvalues for the integer N, including 2, 3, 4, 5, or 6 or more.

Splitting the impedance transformed signal 21 into N split signals canfurther decrease a voltage level of the RF input signal by a factor ofN. Including the splitter transformer 12 can also reduce the impedanceby a factor of N. For example, when the output impedance of the inputimpedance transformer 11 has a value of R_(L), the output impedance ofeach output of the splitter transformer 12 can have a value of R_(L)/N.

As shown in FIG. 3, the RF signal processing circuit 13 can receive thefirst, second, and third split signals 22 a-22 c, and can generatefirst, second, and third processed RF signals 23 a-23 c, respectively.As illustrated in FIG. 3, the RF signal processing circuit 13 includesvariable capacitor arrays 16, which can be used to control a filteringcharacteristic of the RF signal processing circuit 13. The RF signalprocessing circuit 13 further receives a control signal CNTL, which canbe used to control the capacitances of the variable capacitor arrays 16.

The illustrated RF signal processing circuit 13 can be used to processthe split signals 22 a-22 c generated by the splitter transformer 12 togenerate the processed signals 23 a-23 c, respectively. In certainconfigurations, the RF signal processing circuit 13 can includesubstantially identical circuitry in the signal paths between the RFsignal processing circuit's inputs and outputs.

The combiner transformer 14 receives the processed signals 23 a-23 c,which the combiner transformer 14 can combine to generate a combinedsignal 24. The combiner transformer 14 can also provide an impedancetransformation. For example, in a configuration in which each output ofthe RF signal processing circuit 13 has an output impedance of aboutR_(L)/N, the combiner transformer 14 can have an output impedance ofabout R_(L).

The output impedance transformer 15 receives the combined signal 24 fromthe combiner transformer 14, and generates the RF output signal on theRF output OUT. In certain configurations, the combiner transformer 14can have an output impedance R_(L) that is less than 50Ω, and the outputimpedance transformer 15 can be used to provide the RF output signal atan output impedance of about 50Ω.

The illustrated programmable filter 20 provides filtering using the RFsignal processing circuit 13, which processes the split signals 22 a-22c at lower impedance relative to the programmable filter's inputimpedance. Thereafter, the processed signals 23 a-23 c are combined andtransformed up in impedance. For example, in one embodiment, theprogrammable filter's output impedance is about equal to theprogrammable filter's input impedance.

Configuring the programmable filter 20 to process an RF input signal inthis manner can increase the programmable filter's voltage handlingcapability. For example, when the programmable filter 20 has an inputimpedance of about 50Ω, the voltage level of the RF input signal can bedecreased by a factor of about N√{square root over (50/R_(L))} before itis provided to the RF signal processing circuit 13, which may includecircuitry that is sensitive to high voltage conditions. Accordingly, theillustrated programmable filter 20 can be used to process high voltageRF input signals and/or can have enhanced robustness to variations involtage standing wave ratio (VWSR).

Furthermore, configuring the programmable filter 20 to process the RFsignal at lower impedance can enhance the programmable filter'slinearity. In one embodiment, the illustrated configuration can reducethe third-order inter-modulation distortion (IMD3) by a factor of about40 log₁₀ N√{square root over (50/R_(L))} relative to a configuration inwhich an RF input signal is provided directly to an RF signal processingcircuit without impedance transformation or splitting. In oneillustrative example, N can be selected to be equal to 8 and R_(L) canbe selected to be about equal to about 8Ω, and the programmable filtercan provide a linearity improvement of about 52 dB. However, otherconfigurations are possible.

FIG. 4A is a schematic diagram of one embodiment of an RF signalprocessing circuit 30. The RF signal processing circuit 30 includes afirst inductor-capacitor (LC) circuit 31 a, a second LC circuit 31 b, athird LC circuit 31 c, a fourth LC circuit 31 d, a fifth LC circuit 31e, a sixth LC circuit 31 f, a seventh LC circuit 31 g, an eighth LCcircuit 31 h, and a ninth LC circuit 31 i. The RF signal processingcircuit 30 illustrates one embodiment of the RF signal processingcircuit 13 of FIG. 3.

As shown in FIG. 4A, the first, second, and third LC circuits 31 a-31 care arranged in a cascade between a first RF input I₁ and a first RFoutput O₁. Additionally, the fourth, fifth, and sixth LC circuits 31d-31 f are arranged in a cascade between a second RF input I₂ and asecond RF output O₂. Furthermore, the seventh, eighth, and ninth LCcircuits 31 g-31 i are arranged in a cascade between a third RF input I₃and a third RF output O₃.

Although FIG. 4A illustrates a configuration including three RF inputsand three RF outputs, the RF signal processing circuit 30 can be adaptedto include more or fewer inputs and outputs.

The RF signal processing circuit 30 can be used to process RF inputsignals received on the first to third RF inputs I₁-I₃ to generate RFoutput signals on the first to third RF outputs O₁-O₃. As shown in FIG.4A, the RF signal processing circuit 30 receives a control signal CNTL,which can be used to control one or more variable capacitancesassociated with the first to ninth LC circuits 31 a-31 i. By controllingthe LC circuits' capacitances, the control signal CNTL can be used totune a frequency response of the RF signal processing circuit 30.

In one embodiment, the RF signal processing circuit 30 is configured tooperate as a notch filter using techniques including those known in theart, and the control signal CNTL can be used to control a location infrequency of the notch filter's stopband. However, other configurationsare possible.

Although FIG. 4A illustrates a configuration including three LC circuitsarranged in a cascade between each input and output, more or fewer LCcircuits and/or other processing circuitry can be included.

Cascading LC circuits can increase a voltage handling capability of anRF signal processing circuit by limiting a voltage drop acrossindividual circuit components of the LC circuits. For example, incertain implementations, the LC circuits 31 a-31 i are implemented usingMOS capacitors, which can be damaged by large gate-to-drain and/orgate-to-source voltages. By arranging two or more LC circuits in acascade, a voltage drop across the MOS capacitors during operation canbe increased relative to a configuration including a single LC circuitbetween a particular input and output.

The RF signal processing circuit 30 illustrates one embodiment of the RFsignal processing circuit 13 of FIG. 4A. For example, in certainconfigurations, the first to third input RF inputs I₁-I₃ can receive thefirst to third RF split signals 22 a-22 c, respectively, and the firstto third RF outputs O₁-O₃ can generate the first to third processedsignals 23 a-23 c, respectively.

The RF signal processing circuit 30 includes a first signal path betweenthe first RF input I₁ and the first RF output O₁, a second signal pathbetween the second RF input I₂ and the second RF output O₂, and a thirdsignal path between the third RF input I₃ and the third RF output O₃. Incertain configurations, one or more electrical connections can beprovided between corresponding positions along the first to thirdsignals paths. For example, in certain implementations, the RF signalprocessing circuit 30 is used to process substantially identical RFinput signals received on the first to third RF inputs I₁-I₃,respectively, to generate substantially identical RF output signals onthe first to third RF outputs O₁-O₃. In such configurations, electricalconnections can be provided along corresponding positions of signalpaths, since the corresponding positions should have substantially thesame voltage level. Examples of such electrical connections areillustrated in FIG. 4A with dashed lines.

FIG. 4B is a schematic diagram of another embodiment of an RF signalprocessing circuit 40. The RF signal processing circuit 40 includes afirst LC circuit 41 a, a second LC circuit 41 b, a third LC circuit 41c, a fourth LC circuit 41 d, a fifth LC circuit 41 e, a sixth LC circuit41 f, a seventh LC circuit 41 g, an eighth LC circuit 41 h, and a ninthLC circuit 41 i.

The first to ninth LC circuits 41 a-41 i each include an input and anoutput. The first, second, and third LC circuits 41 a-41 c are arrangedin a cascade between the first RF input I1 and the first RF output O1.Additionally, the fourth, fifth, and sixth LC circuits 41 d-41 f arearranged in a cascade between the second RF input I2 and second RFoutput O2. Furthermore, the seventh, eighth, and ninth LC circuits arearranged in a cascade between the third RF input I3 and the third RFoutput O3.

The first LC circuit 41 a includes a first variable capacitor 43 a, asecond variable capacitor 44 a, a first inductor 45 a, a second inductor46 a, and a third inductor 47 a. The first variable capacitor 43 aincludes a first end electrically connected to the input of first LCcircuit 41 a, and a second end electrically connected to a first end ofthe first inductor 45 a. The first inductor 45 a further includes asecond end electrically connected to a first end of the second inductor46 a and to a first end of the third inductor 47 a. The second variablecapacitor 44 a includes a first end electrically connected to a secondend of the second inductor 46 a and a second end electrically connectedto a first voltage V1, which can be, for example, a ground or power lowsupply. The third inductor 47 a further includes a second endelectrically connected to an output of the first LC circuit 41 a.

The second to ninth LC circuits 41 b-41 i include first variablecapacitors 43 b 43 i, second variable capacitors 44 b-44 i, firstinductors 45 b-45 i, second inductors 46 b 46 i, and third inductors 47b-47 i, respectively. Additional details of the second to ninth LCcircuits 41 b 41 i can be similar to those described above with respectto the first LC circuit 41 a.

The control signal CNTL can be used to control variable capacitances ofthe variable capacitors of the first to ninth LC circuits 41 a 41 i,thereby controlling a passband of the RF signal processing circuit 40.In certain implementations, an inductance of the first to ninth LCcircuits 41 a 41 i is substantially fixed or constant.

In certain configurations, all or part of the variable capacitors of anRF signal processing circuit are implemented using variable capacitorarrays fabricated on one or more integrated circuits. For example, asshown in FIG. 4B, in one embodiment, the first variable capacitor 43 a,the fourth variable capacitor 43 d, and the seventh variable capacitor44 g are fabricated as three variable capacitor arrays on a first IC 50.Additionally, the other variable capacitors shown in FIG. 4B can befabricated as variable capacitor arrays on the first IC 50 or on one ormore additional ICs. Although one example of implementing variablecapacitors as variable capacitor arrays has been described, otherconfigurations are possible.

In one embodiment, the control signal CNTL is received over aninterface, such as a serial peripheral interface (SPI) or MobileIndustry Processor Interface radio frequency front end (MIPI RFFE)interface.

As described above, various embodiments of a tunable phasing networkinclude one or more metal oxide semiconductor (MOS) variable capacitorarrays. For various embodiments, a variable capacitor array includes aplurality of variable capacitor cells electrically connected inparallel. Each of the variable capacitor cells can include a cascade oftwo or more pairs of anti-series metal oxide semiconductor (MOS)capacitors between an RF input and an RF output. The pairs ofanti-series MOS capacitors include a first MOS capacitor and a secondMOS capacitor electrically connected in anti-series. A bias voltagegeneration circuit generates bias voltages for biasing the MOScapacitors of the MOS variable capacitor cells.

A MOS capacitor, according to various embodiments, includes a gate thatoperates as an anode, and a source and drain that are electricallyconnected to one another and operate as a cathode. Additionally, a DCbias voltage between the MOS capacitor's anode and cathode can be usedto control the MOS capacitor's capacitance. In certain configurations,two or more pairs of anti-series MOS capacitors are cascaded to operateas a variable capacitor cell. As used herein, a pair of MOS capacitorscan be electrically connected in anti-series or inverse series when thepair of MOS capacitors is electrically connected in series with thefirst and second MOS capacitors' anodes electrically connected to oneanother or with the first and second MOS capacitors' cathodeselectrically connected to one another.

The variable capacitor arrays disclosed herein can exhibit high RFsignal handling and/or power handling capabilities. For example,including two or more pairs of anti-series MOS capacitors in a cascadecan facilitate handling of RF signals with relatively large peak-to-peakvoltage swings by distributing the RF signal voltage across multiple MOScapacitors. Thus, the variable capacitor array can handle RF signals oflarge voltage amplitude and/or high power without overvoltage conditionsthat may otherwise cause transistor damage, such as gate oxide punchthrough.

In certain configurations, the bias voltage generation circuit can biasthe MOS capacitors of a particular variable capacitor cell at a voltagelevel selected from a discrete number of two or more bias voltage levelsassociated with high linearity. Thus, rather than biasing the MOScapacitors at a bias voltage level selected from a continuous tuningvoltage range, the bias voltage generation circuit generates the MOScapacitors' bias voltages by selecting a particular cell's bias voltagelevel from a discrete set of bias voltage levels associated with highlinearity. In one embodiment, the bias voltage generation circuit biasesa particular MOS capacitor either at a first bias voltage levelassociated with an accumulation mode of the MOS capacitor or at a secondbias voltage level associated an inversion mode of the MOS capacitor.

As used herein and as persons having ordinary skill in the art willappreciate, the terms MOS capacitors refer to any types of capacitorsmade from transistors with insulated gates. These MOS capacitors canhave gates made from metals, such as aluminum, and dielectric regionsmade out of silicon oxide. However, these MOS capacitors canalternatively have gates made out of materials that are not metals, suchas poly silicon, and can have dielectric regions implemented not justwith silicon oxide, but with other dielectrics, such as high-kdielectrics. In certain embodiments, the MOS capacitors are implementedusing fabricated using silicon on insulator (SOI) processes. Forexample, an integrated circuit can include a support substrate, a buriedoxide (BOX) layer over the support substrate, and a device layer overthe BOX layer, and the MOS capacitors can be fabricated in the devicelayer.

In certain embodiments, a variable capacitor array omits any switches inthe signal path between the variable capacitor array's RF input and RFoutput. Switches can introduce insertion loss, degrade Q-factor, and/ordecrease linearity. Thus, rather than providing capacitance tuning byopening and closing switches to set a number of active capacitors from acapacitor bank, capacitance tuning can be provided by biasing MOScapacitors of the variable capacitor cells at different bias voltagelevels to provide a desired overall capacitance of the variablecapacitor array. In certain configurations, the variable capacitor cellsof the variable capacitor array can have the same or different weightsor sizes, and the variable capacitor array's overall capacitance isbased on a linear combination of the capacitances of the variablecapacitor cells.

The variable capacitor arrays herein can have high RF voltage handlingcapability, while having a relatively small size, a relatively highQ-factor, a relatively high linearity, and/or a relatively low insertionloss. Furthermore, in certain implementations, a variable capacitorarray can provide sufficient tuning range to provide filtering across avariety of different frequency bands. Accordingly, the variablecapacitor array may be used to provide frequency tuning in a wide rangeof RF electronics, including, for example, programmable filters,programmable resonators, programmable antenna tuners, programmableimpedance matching networks, programmable phase shifters, and/orprogrammable duplexers.

A wireless device such as a smart phone, tablet, or laptop computer cancommunicate over multiple frequency bands using one or more common orshared antennas. A desire to transmit at wider bandwidth and/or overdifferent communications networks has increased a demand for the numberof bands that a wireless device can communicate over. For example, awireless device may be specified to operate using one or more of avariety of communications standards including, for example, GSM/EDGE,IMT-2000 (3G), 4G, Long Term Evolution (LTE), Advanced LTE, IEEE 802.11(Wi-Fi), Mobile WiMAX, Near Field Communication (NFC), GlobalPositioning System (GPS), GLONASS, Galileo, Bluetooth, and the like.Proprietary standards can also be applicable. The complexities ofmulti-band communication can be further exacerbated in configurations inwhich the wireless device is specified to use carrier aggregation.

The metal oxide semiconductor (MOS) capacitors, which can offer enhancedperformance over certain other tunable capacitance structures. Forinstance, certain microelectromechanical systems (MEMS) capacitors canexhibit low Q-factor, poor reliability, and/or limited tuning range.Additionally, other approaches such as coupled resonators can sufferfrom large size and/or cost, and thus can be unsuitable for certainapplications, including smart phones.

FIG. 5 is a schematic diagram of an integrated circuit (IC) 60 accordingto one embodiment. The IC 60 includes a first variable capacitor array61, a second variable capacitor array 62, a third variable capacitorarray 63, and a bias voltage generation circuit 64. The IC 60 includes afirst RF input RF_(IN1), a second RF input RF_(IN2), a third RF inputRF_(IN3), a first RF output RF_(OUT1), a second RF output RF_(OUT2), anda third RF output RF_(OUT3).

The first variable capacitor array 61 includes a first variablecapacitor cell 71 a, a second variable capacitor cell 71 b, and a thirdvariable capacitor cell 71 c. The first to third capacitors cells 71a-71 c are electrically connected in parallel between the first RF inputUm and the first RF output RF_(OUT1). The second variable capacitorarray 62 includes a first variable capacitor cell 72 a, a secondvariable capacitor cell 72 b, and a third variable capacitor cell 72 c.The first to third capacitors cells 72 a 72 c are electrically connectedin parallel between the second RF input RF_(IN2) and the second RFoutput RF_(OUT2). The third variable capacitor array 63 includes a firstvariable capacitor cell 73 a, a second variable capacitor cell 73 b, anda third variable capacitor cell 73 c. The first to third capacitorscells 73 a 73 c are electrically connected in parallel between the thirdRF input RF_(IN3) and the third RF output RF_(OUT3).

Although FIG. 5 illustrates the IC 60 as including three variablecapacitor arrays, the IC 60 can be adapted to include more or fewervariable capacitor arrays. In one embodiment, the IC 60 can includebetween about 4 and about 16 variable capacitor arrays. In anotherembodiment, the IC 60 includes between about 1 and about 3 variablecapacitor arrays. However, other configurations are possible.

Additionally, although FIG. 5 illustrates each variable capacitor arrayas including three variable capacitor cells, the variable capacitorarrays can be adapted to include more or fewer variable capacitor cells.In one embodiment, the IC 60 includes between about 6 and about 12variable capacitor cells. However, a variable capacitor array can beadapted to include other numbers of variable capacitor cells.

The bias voltage generation circuit 64 receives the control signal CNTL,and generates a first bias voltage V_(BIAS1), a second bias voltageV_(BIAS2), and a third bias voltage V_(BIAS3). As shown in FIG. 5, thefirst bias voltage V_(BIAS1) is provided to the first variable capacitorcell 71 a of the first variable capacitor array 61, to the firstvariable capacitor cell 72 a of the second variable capacitor array 62,and to the first variable capacitor cell 73 a of the third variablecapacitor array 63. Additionally, the second bias voltage V_(BIAS2) isprovided to the second variable capacitor cell 71 b of the firstvariable capacitor array 61, to the second variable capacitor cell 72 bof the second variable capacitor array 62, and to the second variablecapacitor cell 73 b of the third variable capacitor array 63.Furthermore, the third bias voltage V_(BIAS3) is provided to the thirdvariable capacitor cell 71 c of the first variable capacitor array 61,to the third variable capacitor cell 72 c of the second variablecapacitor array 62, and to the third variable capacitor cell 73 c of thethird variable capacitor array 63.

The bias voltage generation circuit 64 can be used to control thevoltage levels of the first, second, and third bias voltagesV_(BIAS1)-V_(BIAS3) to control the capacitances of the first to thirdvariable capacitor arrays 61-63.

The illustrated variable capacitor cells can be implemented using MOScapacitors. For example, in certain configurations, two or more pairs ofanti-series MOS capacitors are cascaded to operate as a variablecapacitor cell. Additionally, the first to third bias voltagesV_(BIAS1)-V_(BIAS3) can be used to bias the MOS capacitors at two ormore bias voltages associated with a small amount of capacitancevariation, and thus with high linearity. For example, in one embodiment,the first to third bias voltages V_(BIAS1) V_(BIAS3) can be selectivelycontrolled to bias the MOS capacitors in accumulation or inversion tocontrol the overall capacitance of the arrays.

In certain configurations, the MOS capacitors can be fabricated usingsilicon on insulator (SOI) processes. However, other configurations arepossible, including, for example, implementations in which the MOScapacitors are fabricated using deep sub-micron (DSM) complementarymetal oxide semiconductor (CMOS) processes.

In certain configurations herein, a variable capacitor cell can includepairs of MOS capacitors implemented using anti-series configurations.Configuring a variable capacitor cell in this manner can help cancel thesecond-order intermodulation tones (IM2) and/or control the variation inthe cell's capacitance in the presence of RF signals.

As shown in FIG. 5, the bias voltage generation circuit 64 receives thecontrol signal CNTL, which can be used to select the voltage levels ofthe first, second, and third bias voltages V_(BIAS1)-V_(BIAS3). Incertain configurations, each of the variable capacitor arrays 61-63includes weighted banks of capacitors cells. For example, in oneembodiment, the first variable capacitor cell 71 a, the second variablecapacitor cell 71 b, and the third variable capacitor cell 71 c havedifferent capacitance weights or sizes. For example, the variablecapacitor cells of a particular variable capacitor array can increase insize by a scaling factor, such as 2.

The IC 60 includes a first signal path from the first RF input RF_(IN1)to the first RF output RF_(OUT1) through the first variable capacitorarray 61. Additionally, the IC 60 includes a second signal path from thesecond RF input RF_(IN2) to the second RF output RF_(OUT2) through thesecond variable capacitor array 62, and a third signal path from thethird RF input RF_(IN3) to the third RF output RF_(OUT3) through thethird variable capacitor array 63.

In certain embodiments, the IC 60 does not include any switches in thesignal paths between the IC's inputs and outputs through the variablecapacitor arrays. By configuring the variable capacitor arrays in thismanner, the variable capacitor arrays can have lower insertion lossand/or higher linearity relative to a configuration in which capacitanceis provided by selecting discrete capacitors via switches.

As shown in FIG. 5, multiple variable capacitor arrays can be fabricatedon a common IC, and can share control signals but receive different RFsignals. However, other configurations are possible, such asimplementations in which the variable capacitor arrays receive separatecontrol signals.

FIGS. 6A and 6B are graphs of two examples of capacitance versus biasvoltage. FIG. 6A includes a first graph 91 of capacitance versusvoltage, and FIG. 6B includes a second graph 92 of capacitance versusvoltage.

The first graph 91 includes a high frequency capacitance-voltage (CV)plot 93 for one example of an n-type MOS capacitor. As shown in the CVplot 93, the capacitance of the MOS capacitor can increase with biasvoltage level. The increase in capacitance can be associated with theMOS capacitor transitioning between operating regions or modes. Forexample, at low bias voltage levels, the MOS capacitor can operate in anaccumulation mode in which a majority carrier concentration near thegate dielectric/semiconductor interface is greater than a backgroundmajority carrier concentration of the semiconductor. Additionally, asthe voltage level of the bias voltage increases, the MOS capacitor cantransition from the accumulation mode to a depletion mode in whichminority and majority carrier concentrations near the gatedielectric/semiconductor interface are less than the background majoritycarrier concentration. Furthermore, as the voltage level of the biasvoltage further increases, the MOS capacitor can transition from thedepletion mode to an inversion mode in which the minority carrierconcentration near the gate dielectric/semiconductor interface isgreater than the background majority carrier concentration.

The first graph 91 has been annotated to include an AC signal component94 when biasing the MOS capacitor at a bias voltage level VB. When theAC signal component 94 is not present, the MOS capacitor can have acapacitance C. However, as shown by in FIG. 6A, the AC signal component94 can generate a capacitance variation 95. The capacitance variation 95can be associated with a capacitance variation generated by the ACsignal component 94.

With reference to FIG. 6B, the second graph 92 includes the CV plot 93,which can be as described above. The second graph 92 has been annotatedto include a first AC signal component 96 associated with biasing theMOS capacitor at a first bias voltage level V_(B1), and a second ACsignal component 97 associated with biasing the MOS capacitor at asecond bias voltage level V_(B2).

As shown in FIG. 6B, the first AC signal component 96 can generate afirst capacitance variation 98, and the second AC signal component 97can generate a second capacitance variation 99.

When biased at the first bias voltage level V_(B1) or the second biasvoltage level V_(B2), the MOS capacitor can nevertheless have acapacitance that varies in the presence of AC signals. However, thefirst and second bias voltage levels V_(B1), V_(B2) can be associatedwith DC bias points of the MOS capacitor having relatively smallcapacitance variation or change.

Accordingly, in contrast to the capacitance variation 95 of FIG. 6Awhich has a relatively large magnitude, the first and second capacitancevariations 98, 99 of FIG. 6B have a relatively small magnitude.

In certain embodiments herein, a variable capacitor array includes MOScapacitors that are biased at bias voltages associated with smallcapacitance variation. By biasing the MOS capacitors in this manner, avariable capacitor array can exhibit high linearity.

Such a variable capacitor array can also have less capacitance variationwhen operated in a system using multiple frequency bands. For example,when included in a tunable filter, such as a tunable notch filter and atunable mirror filter, or a tunable matching network, the variablecapacitor array can provide relatively constant capacitance even whentuned to frequency bands that are separated by a wide frequency.

In certain embodiments, the first bias voltage level V_(B1) is selectedto operate in the MOS capacitor in an accumulation mode, and the secondbias voltage level V_(B2) is selected to operate the MOS capacitor in aninversion mode. In certain configurations, biasing a MOS capacitor inthis manner can achieve a capacitance tuning range of 3:1 or more.However, other tuning ranges can be realized, including, for example, atuning range associated with a particular manufacturing process used tofabricate the MOS capacitor.

FIG. 7 is a schematic diagram of an IC 100 according to anotherembodiment. The IC 100 includes a variable capacitor array 101 and abias voltage generation circuit 104. Although FIG. 7 illustrates aconfiguration in which the IC 100 includes one variable capacitor array,the IC 100 can be adapted to include additional variable capacitorarrays and/or other circuitry.

The variable capacitor array 101 includes a first variable capacitorcell 111 a, a second variable capacitor cell 111 b, and a third variablecapacitor cell 111 c, which have been electrically connected in parallelbetween an RF input RF_(IN) and an RF output RF_(OUT). Although theillustrated variable capacitor array 101 includes three variablecapacitor cells, the variable capacitor array 101 can be adapted toinclude more or fewer variable capacitor cells.

The bias voltage generation circuit 104 receives the control signalCNTL, and generates a first bias voltage 105 a for the first variablecapacitor cell 111 a, a second bias voltage 105 b for the secondvariable capacitor cell 111 b, and a third bias voltage 105 c for thethird variable capacitor cell 111 c.

In the illustrated configuration, the control signal CNTL can be used toset the voltage level of the first bias voltage 105 a to a first biasvoltage level V_(B1) or to a second bias voltage level V_(B2).Similarly, the control signal CNTL can be used to set the voltage levelof the second bias voltage 105 b to the first bias voltage level V_(B1)or to the second bias voltage level V_(B2), and to set the voltage levelof the third bias voltage 105 c to the first bias voltage level V_(B1)or to the second bias voltage level V_(B2).

By controlling the voltage levels of the bias voltages to the first orsecond bias voltage levels V_(B1), V_(B2), the variable capacitor array101 can exhibit a small variation in capacitance in the presence of anRF signal at the RF input RF_(IN). Accordingly, the variable capacitorarray 101 can exhibit high linearity in the presence of RF signals.

The control signal CNTL can control an overall capacitance of thevariable capacitor array 101. For example, the size of the first,second, and third MOS capacitors cells 111 a 111 c can be weightedrelative to one another, and an overall capacitance of the variablecapacitor array 101 can be based on a sum of the capacitances of thearray's variable capacitor cells.

In one embodiment, the variable capacitor array's variable capacitorcells are scaled by a factor of 2, and each of the variable capacitorcells includes k pairs of anti-series MOS capacitors connected in acascade. For example, a second variable capacitor cell of the variablecapacitor array can have a size that is about a factor of 2 relative toa first variable capacitor cell of the variable capacitor array.Additionally, an nth variable capacitor cell in the array can have asize that is about 2^(n-1) that of the first variable capacitor cell,where n is an integer greater than or equal to 2. Although one possiblevariable capacitor array sizing scheme has been described, otherconfigurations are possible.

When a variable capacitor array includes n variable capacitor cells thatare scaled by a factor of 2 relative to one another and that include kpairs of anti-series MOS capacitors in a cascade, the bias voltagegeneration circuit 104 can control the array's first variable capacitorcell to a capacitance of C₁/2k or C₂/2k by biasing the first variablecapacitor cell with the first bias voltage level V_(B1) or the secondbias voltage level V_(B2). Additionally, the bias voltage generationcircuit 104 can control the array's second variable capacitor cell to acapacitance of 2¹*C₁/2k or 2¹*C₂/2k by biasing the second variablecapacitor cell with the first bias voltage level V_(B1) or the secondbias voltage level VB2. Furthermore, the bias voltage generation circuit104 can control the array's nth variable capacitor cell to a capacitanceof 2^(n-1)*C₁/2k or 2^(n-1)*C₂/2k by biasing the nth variable capacitorcell with the first bias voltage level V_(B1) or the second bias voltagelevel V_(B2).

Configuring the bias voltage generation circuit 104 to control a biasvoltage to one of two voltage levels can simplify a coding schemeassociated with the control signal CNTL. For example, in such aconfiguration, the control signal CNTL can comprise a digital controlsignal, and individual bits of the digital control signal can be used tocontrol the array's bias voltages to a particular bias voltage level.Although one possible coding scheme of the control signal CNTL has beendescribed, other configurations are possible.

FIG. 8 is a schematic diagram of an IC 120 according to anotherembodiment. The IC 120 includes a variable capacitor array 121 and abias voltage generation circuit 124. Although FIG. 8 illustrates aconfiguration in which the IC 120 includes one variable capacitor array,the IC 100 can be adapted to include additional variable capacitorarrays and/or other circuitry.

The variable capacitor array 121 includes a first variable capacitorcell 121 a, a second variable capacitor cell 121 b, and a third variablecapacitor cell 121 c, which have been electrically connected in parallelbetween an RF input RF_(IN) and an RF output RF_(OUT). The firstvariable capacitor cell 121 a includes a cascade of a first pair ofanti-series MOS capacitors 141 a, a second pair of anti-series MOScapacitors 141 b, and a third pair of anti-series MOS capacitors 141 c.The second variable capacitor cell 121 b includes a cascade of a firstpair of anti-series MOS capacitors 142 a, a second pair of anti-seriesMOS capacitors 142 b, and a third pair of anti-series MOS capacitors 142c. The third variable capacitor cell 121 c includes a cascade of a firstpair of anti-series MOS capacitors 143 a, a second pair of anti-seriesMOS capacitors 143 b, and a third pair of anti-series MOS capacitors 143c. Although the illustrated variable capacitor array 121 includes threevariable capacitor cells, the variable capacitor array 121 can beadapted to include more or fewer variable capacitor cells. Additionally,although the illustrated variable capacitor cells each include a cascadeof three pairs of anti-series MOS capacitors, the variable capacitorcells can include more or fewer pairs of anti-series MOS capacitors.

The bias voltage generation circuit 124 receives the control signalCNTL, and generates a first bias voltage V_(BIAS1) for the firstvariable capacitor cell 131 a, a second bias voltage V_(BIAS2) for thesecond variable capacitor cell 131 b, and a third bias voltage V_(BIAS3)for the third variable capacitor cell 131 c. In certain configurations,the bias voltage generation circuit 124 can also be used to generate abody bias voltage V_(BODY), which can be used to control the bodyvoltages of MOS capacitors of the variable capacitor array 121.

Additional details of the integrated circuit 120 can be similar to thosedescribed earlier.

FIG. 9A is a schematic diagram of a variable capacitor cell 150according to one embodiment. The variable capacitor cell 150 includes afirst pair of anti-series MOS capacitors 151, a second pair ofanti-series MOS capacitors 152, a third pair of anti-series MOScapacitors 153, a first DC biasing resistor 171, a second DC biasingresistor 172, a third DC biasing resistor 173, a fourth DC biasingresistor 174, a first control biasing resistor 181, a second controlbiasing resistor 182, and a third control biasing resistor 183.

Although the variable capacitor cell 150 is illustrated as includingthree pairs of anti-series MOS capacitors, the teachings herein areapplicable to configurations including more or fewer pairs ofanti-series MOS capacitors. For example, in one embodiment, a variablecapacitor cell includes a cascade of between 2 and 18 pairs ofanti-series MOS capacitors.

In the illustrated configuration, each of the pairs of anti-series MOScapacitors 151-153 includes two MOS capacitors electrically connected inanti-series or inverse series. For example, the first pair ofanti-series MOS capacitors 151 includes a first MOS capacitor 161 and asecond MOS capacitor 162. The first and second MOS capacitors 161, 162have anodes associated with transistor gates and cathodes associatedwith transistor source and drain regions. As shown in FIG. 9A, the anodeof the first MOS capacitor 161 is electrically connected to the anode ofthe second MOS capacitor 162. Additionally, the second pair ofanti-series MOS capacitors 152 includes a third MOS capacitor 163 and afourth MOS capacitor 164, and the anode of the third MOS capacitor 163is electrically connected to the anode of the fourth MOS capacitor 164.Furthermore, the third pair of anti-series MOS capacitors 153 includesfifth MOS capacitor 165 and a sixth MOS capacitor 166, and the anode ofthe fifth MOS capacitor 165 is electrically connected to the anode ofthe sixth MOS capacitor 166.

As shown in FIG. 9A, the first to third pairs of anti-series MOScapacitors 151-153 are arranged in a cascade between the RF inputRF_(IN) and the RF output RF_(OUT). For example, the cathode of thefirst MOS capacitor 161 is electrically connected to the RF inputRF_(IN), and the cathode of the second MOS capacitor 162 is electricallyconnected to the cathode of the third MOS capacitor 163. Additionally,the cathode of the fourth MOS capacitor 164 is electrically connected tothe cathode of the fifth MOS capacitor 165, and a cathode of the sixthMOS capacitor 166 is electrically connected to the RF output RF_(OUT).

Arranging two or more pairs of anti-series MOS capacitors in a cascadecan increase a voltage handling capability of a variable capacitor cellrelative to a configuration including a single pair of anti-series MOScapacitors. For example, arranging two or more pairs of anti-series MOScapacitors in a cascade can increase a voltage handling and/or powerhandling capability of the variable capacitor cell by distributing RFsignal voltage across multiple MOS capacitors.

Accordingly, cascading several pairs of anti-series MOS capacitors canachieve high voltage operation of a variable capacitor cell.

Additionally, the illustrated variable capacitor cell 150 includes pairsof MOS capacitors that are electrically connected in anti-series, whichcan decrease capacitance variation in the presence of RF signals. Forexample, when the first and second variable capacitors are each biasedwith a particular bias voltage, the variable capacitors' capacitance maychange when an RF input signal is received on the RF input RF_(IN).However, a capacitance variation ΔC between MOS capacitors in a givenpair can have about equal magnitude, but opposite polarity.

For instance, in the presence of an RF input signal that generates acapacitance variation having a magnitude ΔC, a first MOS capacitor of apair of anti-series MOS capacitors may have a capacitance C_(V)+ΔC,while the second MOS capacitor may have a capacitance C_(V)−ΔC. Thus,the total capacitance of the anti-series combination of the first andsecond MOS capacitors 121, 122 can be about equal to ½C_(V)½ΔC²/C_(V).Since ½ΔC² is typically much smaller than ΔC, the anti-series MOScapacitors can exhibit small capacitance variation when RF signalspropagate through the variable capacitor cell.

Accordingly, the illustrated variable capacitor cell 150 can providereduced capacitance variation in the presence of RF signals.

In the illustrated configuration, the first to fourth DC biasingresistors 171-174 have been used to bias the cathodes of the MOScapacitors 161-166 with the first voltage V₁, which can be a ground,power low supply, or other reference voltage in certain implementations.Additionally, the first to third control biasing resistors 181-183 areused to bias the anodes of the MOS capacitors 161-166 with the biasvoltage V_(BIAS).

In one embodiment, the DC biasing resistors 171-174 have a resistanceselected in the range of 10 kΩ to 10,000 kΩ, and the control biasingresistors 181-183 have a resistance selected in the range of 10 kΩ to10,000 kΩ. Although one example of resistance values have been provided,other configurations are possible. For example, choosing relatively lowresistance values for the biasing resistors can increase control over DCbiasing conditions, but can also undesirably increase signal loss and/ordegrade linearity since the resistors operate in shunt to an RF signalpropagating through the variable capacitor cell. Accordingly, resistancevalues can vary depending on application, fabrication process, and/ordesired performance specifications.

The bias voltages across the MOS capacitors 161-166 can be based on avoltage difference between the bias voltage V_(BIAS) and the firstvoltage V₁. Additionally, a bias voltage generation circuit, such as thebias voltage generation circuit 64 of FIG. 5, can be used to control avoltage level of the bias voltage V_(BIAS) to control the variablecapacitor cell's capacitance between the RF input RF_(IN) and the RFoutput RF_(OUT).

In certain configurations, the bias voltage generation circuit cancontrol the bias voltage V_(BIAS) to a voltage level selected from adiscrete number of two or more bias voltage levels associated with highlinearity. Thus, rather than biasing the MOS capacitors at a biasvoltage level selected from a continuous tuning voltage range, the biasvoltage generation circuit generates the MOS capacitors' bias voltagesby selecting a particular cell's bias voltage level from a discrete setof bias voltage levels associated with high linearity. In oneembodiment, the bias voltage generation circuit biases a particular MOScapacitor either at a first bias voltage level associated with anaccumulation mode of the MOS capacitor or at a second bias voltage levelassociated an inversion mode of the MOS capacitor.

Biasing the MOS capacitors 161-166 in this manner can improve linearityrelative to a configuration in which the MOS capacitors 161-166 arebiased at a bias voltage level selected from a continuous tuning voltagerange. For example, a MOS capacitor can exhibit a change in capacitancein response to changes in an applied RF signal, and a magnitude of thecapacitance change can vary with the MOS capacitor's bias voltage level.

Accordingly, the illustrated variable capacitor cell 150 can providehigh linearity between the RF input RF_(IN) and the RF output RF_(OUT).

FIG. 9B is a circuit diagram of a variable capacitor cell 160 accordingto one embodiment. The variable capacitor cell 160 includes a first pairof anti-series MOS capacitors 191, a second pair of anti-series MOScapacitors 192, a third pair of anti-series MOS capacitors 193, a firstDC biasing resistor 171, a second DC biasing resistor 172, a third DCbiasing resistor 173, a fourth DC biasing resistor 174, a first controlbiasing resistor 181, a second control biasing resistor 182, and a thirdcontrol biasing resistor 183. Although the variable capacitor cell 160is illustrated as including three pairs of anti-series MOS capacitors,the teachings herein are applicable to configurations including more orfewer pairs of anti-series MOS capacitors.

The variable capacitor cell 160 of FIG. 9B is similar to the variablecapacitor cell 150 of FIG. 9A, except that the variable capacitor cell160 illustrates a different anti-series configuration of the pairs ofanti-series MOS capacitors 191-193.

In particular, in contrast to the variable capacitor cell 150 of FIG. 9Ain which the anodes of the MOS capacitors of a given pair areelectrically connected to one another, the variable capacitor cell 160of FIG. 9B illustrates a configuration in which the cathodes of a givenpair of MOS capacitors are electrically connected to one another. Forexample, the first pair of MOS capacitors 191 includes a first MOScapacitor 201 and a second MOS capacitor 202, and the cathodes of thefirst and second MOS capacitors 201, 202 are electrically connected toone another. Similarly, the second pair of MOS capacitors 192 includes athird MOS capacitor 203 and a fourth MOS capacitor 204, and the cathodesof the third and fourth MOS capacitors 203, 204 are electricallyconnected to one another. Likewise, the third pair of MOS capacitors 193includes a fifth MOS capacitor 205 and a sixth MOS capacitor 206, andthe cathodes of the fifth and sixth MOS capacitors 205, 206 areelectrically connected to one another.

As shown in FIG. 9B, the pairs of anti-series MOS capacitors 191-193 areelectrically connected in a cascade between the RF input RF_(IN) and theRF output RF_(OUT). For example, the anode of the first MOS capacitor201 is electrically connected to the RF input RF_(IN), and the anode ofthe second MOS capacitor 202 is electrically connected to the anode ofthe third MOS capacitor 203. Additionally, the anode of the fourth MOScapacitor 204 is electrically connected to the anode of the fifth MOScapacitor 205, and an anode of the sixth MOS capacitor 206 iselectrically connected to the RF output RF_(OUT).

In the illustrated configuration, the first to fourth DC biasingresistors 171 174 are used to bias the anodes of the MOS capacitors201-206 with the first voltage V₁, which can be a ground, power lowsupply, or other reference voltage in certain implementations.Additionally, the first to third control biasing resistors 181-183 areused to bias the cathodes of the MOS capacitors 201-206 with the biasvoltage V_(BIAS).

In certain configurations, the variable capacitor cell 150 of FIG. 9Acan be more robust against damage from electrostatic discharge (ESD)events relative to the variable capacitor cell 160 of FIG. 9B.

For example, the RF input RF_(IN) and RF output RF_(OUT) of a variablecapacitor cell may be electrically connected to input and output pins ofan IC on which the variable capacitor cell is fabricated. Since a MOScapacitor's source and drain regions typically can withstand a greatervoltage relative to the MOS capacitor's gate region when fabricatedusing certain manufacturing processes, the variable capacitor cell 150of FIG. 9A may exhibit a greater robustness to ESD events or otherovervoltage conditions relative to the variable capacitor cell 160 ofFIG. 9B.

Additional details of the variable capacitor cell 160 can be similar tothose described earlier.

FIG. 10A is a variable capacitor cell 220 according to anotherembodiment. The variable capacitor cell 220 of FIG. 10A is similar tothe variable capacitor cell 150 of FIG. 9A, except that the variablecapacitor cell 220 of FIG. 10A further includes a first diode 221, asecond diode 222, a third diode 223, a fourth diode 224, a fifth diode225, and a sixth diode 226.

As shown in FIG. 10A, the diodes 221-226 are electrically connectedbetween the body and gate of the MOS capacitors 161-166, respectively.In particular, the anodes of the diodes 221-226 are electricallyconnected to the bodies of the MOS capacitors 161-166, respectively, andthe cathodes of the diodes 221-226 are electrically connected to thegates of the MOS capacitors 161-166, respectively. The diodes 221-226can be included in a variety of manufacturing processes, such assilicon-on-insulator (SOI) processes. In certain configurations, thediodes 221-226 are implemented as p-n junction diodes. For example, ann-type MOS capacitor can include a p-type body region, and an n-typeactive region can be included in the p-type body region and electricallyconnected to the gate via metallization to provide a forward p-njunction diode from body to gate.

Including the diodes 221-226 can enhance the performance in the presenceof RF signaling conditions, including, for example, enhanced performancein the presence of voltage changes to an RF signal over a signal cycle.For example, the diodes 221-226 can increase voltage headroom of the MOScapacitors 161-166 relative to a configuration in which the diodes221-226 are omitted. Additionally, the diodes 221-226 can aid in betterdistributing an RF signal voltage across the MOS capacitors 161-166,thereby preventing large voltage build-up across a particular MOScapacitor in the cascade. Thus, the illustrated configuration canexhibit greater signal handling and/or power handling capabilityrelative to a configuration that omits the diodes 221-226.

Additional details of the variable capacitor cell 220 can be similar tothose described earlier.

FIG. 10B is a circuit diagram of a variable capacitor cell 230 accordingto another embodiment. The variable capacitor cell 230 of FIG. 10B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 230 of FIG. 10B further includes the first tosixth diodes 221-226.

As shown in FIG. 10B, the anodes of the diodes 221-226 are electricallyconnected to the bodies of the MOS capacitors 201-206, respectively, andthe cathodes of the diodes 221-226 are electrically connected to thegates of the MOS capacitors 201-206, respectively. Including the diodes221-226 can improve RF signal voltage distribution and/or increasevoltage headroom of the MOS capacitors 201-206.

Additional details of the variable capacitor cell 230 can be similar tothose described earlier.

FIG. 11A is a circuit diagram of a variable capacitor cell 240 accordingto another embodiment. The variable capacitor cell 240 of FIG. 11A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 240 of FIG. 11A further includes a first bodybiasing resistor 241, a second body biasing resistor 242, a third bodybiasing resistor 243, a fourth body biasing resistor 244, a fifth bodybiasing resistor 245, and a sixth body biasing resistor 246.

The body biasing resistor 241-246 are used to bias the bodies of the MOScapacitors 161-166 with a body bias voltage V_(BODY). Including the bodybiasing resistors 241-246 can aid in increasing the voltage headroom ofthe MOS capacitors 161-166 in the presence of RF voltage swing. Incertain configurations, the body bias voltage V_(BODY) is generated by abias voltage generation circuit, such as the bias voltage generationcircuit 124 of FIG. 8.

The body biasing resistors 241-246 can have any suitable resistancevalue. In one embodiment, the body biasing resistors 241-246 have aresistance selected in the range of 10 kΩ to 10,000 kΩ. Although oneexample of resistance values have been provided, other configurationsare possible, such as resistance values selected for a particularapplication, fabrication process, and/or desired performancespecifications.

Additional details of the variable capacitor cell 240 can be similar tothose described earlier.

FIG. 11B is a circuit diagram of a variable capacitor cell 250 accordingto another embodiment. The variable capacitor cell 250 of FIG. 11B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 250 of FIG. 11B further includes the first tosixth body biasing resistors 241 246.

As shown in FIG. 11B, the body biasing resistors 241 246 areelectrically connected between the body bias voltage V_(BODY) and thebodies of the MOS capacitors 201-206, respectively. Including the bodybiasing resistors 241 246 can increase voltage headroom of the MOScapacitors 201 206 in the presence of amplitude change or swing of an RFsignal.

Additional details of the variable capacitor cell 250 can be similar tothose described earlier.

FIG. 12A is a circuit diagram of a variable capacitor cell 260 accordingto another embodiment. The variable capacitor cell 260 of FIG. 12A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 260 of FIG. 12A further includes a first signalswing compensation capacitor 261, a second signal swing compensationcapacitor 262, and a third signal swing compensation capacitor 263.

As shown in FIG. 12A, the first signal swing compensation capacitor 261is electrically connected in parallel with the first pair of anti-seriesMOS capacitors 151. For example, the first signal swing compensationcapacitor 261 includes a first end electrically connected to the cathodeof the first MOS capacitor 161 and a second end electrically connectedto the cathode of the second MOS capacitor 162. Similarly, the secondsignal swing compensation capacitor 262 is electrically connected inparallel with the second pair of anti-series MOS capacitors 152, and thethird signal swing compensation capacitor 263 is electrically connectedin parallel with the third pair of anti-series MOS capacitors 153.

The signal swing compensation capacitors 261-263 can be used to balanceor compensate for differences in voltage, current, and/or phase betweenpairs of anti-series MOS capacitors. Absent compensation, variation involtage, current, and/or phase between MOS capacitors may degrade thevariable capacitor cell's linearity.

In certain configurations, the capacitance values of the signal swingcompensation capacitors 261-263 can be individually selected to improvevoltage, current, and/or phase balancing between MOS capacitors 161-166.For example, even when the MOS capacitors 161-166 are implemented withthe same size and/or geometry, the capacitance values of the signalswitch compensation capacitors 261-263 can be individually selected toprovide improve compensation in the presence of RF signaling conditions.In one embodiment, the first signal swing compensation capacitor 261 hasa capacitance value that is greater than that of the second signal swingcompensation capacitor 262, and the second signal swing compensationcapacitor 262 has a capacitance value that is greater than that of thethird signal swing compensation capacitor 263. Sizing the signal swingcompensation capacitors in this manner may provide enhanced balancing incertain configurations, such as configurations in which large amplitudeRF signals are received at the RF input RF_(IN).

Additional details of the variable capacitor cell 260 can be similar tothose described earlier.

FIG. 12B is a circuit diagram of a variable capacitor cell 270 accordingto another embodiment. The variable capacitor cell 270 of FIG. 12B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 270 of FIG. 12B further includes the signalswing compensation capacitors 261-263.

As shown in FIG. 12B, the first signal swing compensation capacitor 261is electrically connected in parallel with the first pair of anti-seriesMOS capacitors 191. For example, the first signal swing compensationcapacitor 261 includes a first end electrically connected to the anodeof the first MOS capacitor 201 and a second end electrically connectedto the anode of the second MOS capacitor 202. Similarly, the secondsignal swing compensation capacitor 262 is electrically connected inparallel with the second pair of anti-series MOS capacitors 192, and thethird signal swing compensation capacitor 263 is electrically connectedin parallel with the third pair of anti-series MOS capacitors 193.

The signal swing compensation capacitors 261-263 can be included tobalance differences in voltage, current, and/or phase between adjacentMOS capacitors, thereby improving linearity of the variable capacitorcell.

Additional details of the variable capacitor cell 270 can be similar tothose described earlier.

FIG. 13A is a circuit diagram of a variable capacitor cell 280 accordingto another embodiment. The variable capacitor cell 280 of FIG. 13A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 280 of FIG. 13A further includes the diodes221-226 and the signal swing compensation capacitors 261-263.

Additional details of the variable capacitor cell 280 can be similar tothose described earlier.

FIG. 13B is a circuit diagram of a variable capacitor cell 290 accordingto another embodiment. The variable capacitor cell 290 of FIG. 13B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 290 of FIG. 13B further includes the diodes221-226 and the signal swing compensation capacitors 261-263.

Additional details of the variable capacitor cell 290 can be similar tothose described earlier.

FIG. 14A is a circuit diagram of a variable capacitor cell 300 accordingto another embodiment. The variable capacitor cell 300 of FIG. 14A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 300 of FIG. 14A further includes the bodybiasing resistors 241-246 and the signal swing compensation capacitors261-263.

Additional details of the variable capacitor cell 300 can be similar tothose described earlier.

FIG. 14B is a circuit diagram of a variable capacitor cell 310 accordingto another embodiment. The variable capacitor cell 310 of FIG. 14B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 310 of FIG. 14B further includes the bodybiasing resistors 241-246 and the signal swing compensation capacitors261-263.

Additional details of the variable capacitor cell 310 can be similar tothose described earlier.

FIG. 15A is a circuit diagram of a variable capacitor cell 320 accordingto another embodiment. The variable capacitor cell 320 of FIG. 15A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 320 of FIG. 15A further includes a first driftprotection resistor 321, a second drift protection resistor 322, and athird drift protection resistor 323.

As shown in FIG. 15A, the first drift protection resistor 321 iselectrically connected in parallel with the first pair of anti-seriesMOS capacitors 151. For example, the first drift protection resistor 321includes a first end electrically connected to the cathode of the firstMOS capacitor 161 and a second end electrically connected to the cathodeof the second MOS capacitor 162. Similarly, the second drift protectionresistor 322 is electrically connected in parallel with the second pairof anti-series MOS capacitors 152, and the third drift protectionresistor 323 is electrically connected in parallel with the third pairof anti-series MOS capacitors 153.

The drift protection resistor 321-323 can be used to balance DCoperating points across the MOS capacitors 161-166, thereby enhancingperformance in the presence of RF amplitude variation or swing. Asdescribed earlier, a capacitance provided by a MOS capacitor changeswith a voltage difference across the MOS capacitor's anode and cathode.Accordingly, balancing the DC operating point across the MOS capacitors161-166 can help prevent the capacitances values of the MOS capacitors161-166 from drifting and becoming unstable in the presence of RFsignaling conditions.

In one embodiment, the drift protection resistors 321-323 have aresistance selected in the range of 5 kΩ to 1,000 kΩ. Although oneexample of resistance values have been provided, other configurationsare possible. For example, choosing relatively low resistance values forthe drift protection resistors can reduce capacitance value drift due toRF signal swing, but can also impact signaling performance since theresistors are electrically in series between the RF input RF_(IN) andthe RF output RF_(OUT). Accordingly, resistance values can varydepending on application, fabrication process, and/or desiredperformance specifications.

Additional details of the variable capacitor cell 320 can be similar tothose described earlier.

FIG. 15B is a circuit diagram of a variable capacitor cell 330 accordingto another embodiment. The variable capacitor cell 330 of FIG. 15B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 330 of FIG. 15B further includes the driftprotection resistors 321-323.

As shown in FIG. 15B, the first drift protection resistor 321 iselectrically connected in parallel with the first pair of anti-seriesMOS capacitors 191. For example, the first drift protection resistor 321includes a first end electrically connected to the anode of the firstMOS capacitor 201 and a second end electrically connected to the anodeof the second MOS capacitor 202. Similarly, the second drift protectionresistor 322 is electrically connected in parallel with the second pairof anti-series MOS capacitors 192, and the third drift protectionresistor 323 is electrically connected in parallel with the third pairof anti-series MOS capacitors 193.

The drift protection resistors 321-323 can be included to prevent thecapacitances values of the MOS capacitors 201-206 from drifting andbecoming unstable in the presence of RF signaling conditions.

Additional details of the variable capacitor cell 330 can be similar tothose described earlier.

FIG. 16A is a circuit diagram of a variable capacitor cell 340 accordingto another embodiment. The variable capacitor cell 340 of FIG. 16A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 340 of FIG. 16A further includes the diodes221-226 and the drift protection resistors 321-323.

Additional details of the variable capacitor cell 340 can be similar tothose described earlier.

FIG. 16B is a circuit diagram of a variable capacitor cell 350 accordingto another embodiment. The variable capacitor cell 350 of FIG. 16B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 350 of FIG. 16B further includes the diodes221-226 and the drift protection resistors 321-323.

Additional details of the variable capacitor cell 350 can be similar tothose described earlier.

FIG. 17A is a circuit diagram of a variable capacitor cell 360 accordingto another embodiment. The variable capacitor cell 360 of FIG. 17A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 360 of FIG. 17A further includes the bodybiasing resistors 241-246 and the drift protection resistors 321-323.

Additional details of the variable capacitor cell 360 can be similar tothose described earlier.

FIG. 17B is a circuit diagram of a variable capacitor cell 370 accordingto another embodiment. The variable capacitor cell 370 of FIG. 17B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 370 of FIG. 17B further includes the bodybiasing resistors 241-246 and the drift protection resistors 321-323.

Additional details of the variable capacitor cell 370 can be similar tothose described earlier.

FIG. 18A is a circuit diagram of a variable capacitor cell 380 accordingto another embodiment. The variable capacitor cell 380 of FIG. 18A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 380 of FIG. 18A further includes a first feedforward capacitor 381, a second feed-forward capacitor 382, and a thirdfeed forward capacitor 383.

As shown in FIG. 18A, the first feed forward capacitor 381 iselectrically connected between the RF input RF_(IN) and an intermediatenode of the first pair of anti-series MOS capacitors 151. For example,the first feed forward capacitor 381 is electrically connected betweenthe RF input RF_(IN) and the anodes of the first and second MOScapacitors 161, 162. Additionally, the second feed-forward capacitor 382is electrically connected between the intermediate node of the firstpair of anti-series MOS capacitors 151 and an intermediate node of thesecond pair of anti-series MOS capacitors 152. For example, the secondfeed forward capacitor 382 includes a first end electrically connectedto the anodes of the first and second MOS capacitors 161, 162 and asecond end electrically connected to anodes of the third and fourth MOScapacitors 163, 164. Furthermore, the third feed forward capacitor 383is electrically connected between the intermediate node of the secondpair of anti-series MOS capacitors 152 and an intermediate node of thethird pair of anti-series MOS capacitors 153. For example, the thirdfeed forward capacitor 383 includes a first end electrically connectedto the anodes of the third and fourth MOS capacitors 163, 164, and asecond end electrically connected to anodes of the fifth and sixth MOScapacitors 165, 166.

The feed forward capacitors 381-383 can be used to balance or compensatefor differences in voltage, current, and/or phase between MOScapacitors. For example, the feed forward capacitors 381-383 can be usedto balance an RF voltage drop across the MOS capacitors 161-166, therebyimproving the linearity of the variable capacitor cell.

In certain configurations, the feed forward capacitors 381-383 can beindividually selected to improve voltage, current, and/or phasebalancing between MOS capacitors 161-166. For example, even when the MOScapacitors 161-166 are implemented with the same size and/or geometry,the capacitance values of the feed forward capacitors 381-383 can beindividually selected to provide improve compensation in the presence ofRF signaling conditions. In one embodiment, the first feed forwardcapacitor 381 has a capacitance value that is greater than that of thesecond feed forward capacitor 382, and the second feed forward capacitor382 has a capacitance value that is greater than that of the third feedforward capacitor 383. Sizing the feed forward capacitors in this mannermay provide enhanced balancing in certain configurations, such asconfigurations in which large amplitude RF signals are received at theRF input RF_(IN).

Additional details of the variable capacitor cell 380 can be similar tothose described earlier.

FIG. 18B is a circuit diagram of a variable capacitor cell 390 accordingto another embodiment. The variable capacitor cell 390 of FIG. 18B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 390 of FIG. 18B further includes the feedforward capacitors 381-383.

As shown in FIG. 18B, the first feed forward capacitor 381 iselectrically connected between the RF input RF_(IN) and an intermediatenode of the first pair of anti-series MOS capacitors 191. For example,the first feed forward capacitor 381 is electrically connected betweenthe RF input RF_(IN) and the cathodes of the first and second MOScapacitors 201, 202. Additionally, the second feed-forward capacitor 382is electrically connected between the intermediate node of the firstpair of anti-series MOS capacitors 191 and an intermediate node of thesecond pair of anti-series MOS capacitors 192. For example, the secondfeed forward capacitor 382 includes a first end electrically connectedto the cathodes of the first and second MOS capacitors 201, 202 and asecond end electrically connected to cathodes of the third and fourthMOS capacitors 203, 204. Furthermore, the third feed forward capacitor383 is electrically connected between the intermediate node of thesecond pair of anti-series MOS capacitors 192 and an intermediate nodeof the third pair of anti-series MOS capacitors 193. For example, thethird feed forward capacitor 383 includes a first end electricallyconnected to the cathodes of the third and fourth MOS capacitors 203,204, and a second end electrically connected to cathodes of the fifthand sixth MOS capacitors 205, 206.

The feed forward capacitors 381 383 can be included to balancedifferences in voltage, current, and/or phase between MOS capacitors,thereby improving linearity of the variable capacitor cell.

Additional details of the variable capacitor cell 390 can be similar tothose described earlier.

FIG. 19A is a circuit diagram of a variable capacitor cell 400 accordingto another embodiment. The variable capacitor cell 400 of FIG. 19A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 400 of FIG. 19A further includes the diodes221-226 and the feed forward capacitors 381-383.

Additional details of the variable capacitor cell 400 can be similar tothose described earlier.

FIG. 19B is a circuit diagram of a variable capacitor cell 410 accordingto another embodiment. The variable capacitor cell 410 of FIG. 19B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 410 of FIG. 19B further includes the diodes221-226 and the feed forward capacitors 381-383.

Additional details of the variable capacitor cell 410 can be similar tothose described earlier.

FIG. 20A is a circuit diagram of a variable capacitor cell 420 accordingto another embodiment. The variable capacitor cell 420 of FIG. 20A issimilar to the variable capacitor cell 150 of FIG. 9A, except that thevariable capacitor cell 420 of FIG. 20A further includes the bodybiasing resistors 241-246 and the feed forward capacitors 381-383.

Additional details of the variable capacitor cell 420 can be similar tothose described earlier.

FIG. 20B is a circuit diagram of a variable capacitor cell 430 accordingto another embodiment. The variable capacitor cell 430 of FIG. 20B issimilar to the variable capacitor cell 160 of FIG. 9B, except that thevariable capacitor cell 430 of FIG. 20B further includes the bodybiasing resistors 241-246 and the feed forward capacitors 381-383.

Additional details of the variable capacitor cell 430 can be similar tothose described earlier.

FIG. 21A is a circuit diagram of a variable capacitor cell 440 accordingto another embodiment. The variable capacitor cell 440 of FIG. 21A issimilar to the variable capacitor cell 320 of FIG. 15A, except that thevariable capacitor cell 440 of FIG. 21A omits the first to fourth DCbiasing resistors 171-174.

As described earlier, the drift protection resistor 321-323 can be usedto balance DC operating points across the MOS capacitors 161-166,thereby enhancing performance in the presence of RF amplitude variationor swing. In the illustrated configuration, the first to fourth DCbiasing resistors 171-174 have been omitted in favor of controlling theDC bias voltage at the cathodes of the MOS capacitors 161-166 using thedrift protection resistors 321-323. For example, in the illustratedconfiguration, the DC bias voltage at the cathodes of the MOS capacitors161-166 can be controlled to a DC bias voltage of the RF input RF_(IN)and RF output RF_(OUT). Additionally, one of the terminals RF_(IN) orRF_(OUT) may be grounded when used in a shunt configuration, thuseliminating the need of first to fourth DC biasing resistors 171-174.

Additional details of the variable capacitor cell 440 can be similar tothose described earlier.

FIG. 21B is a circuit diagram of a variable capacitor cell 450 accordingto another embodiment. The variable capacitor cell 450 of FIG. 21B issimilar to the variable capacitor cell 330 of FIG. 15B, except that thevariable capacitor cell 450 of FIG. 21B omits the first to fourth DCbiasing resistors 171-174.

As shown in FIG. 21B, the first to fourth DC biasing resistors 171-174have been omitted in favor of controlling the DC bias voltage at theanodes of the MOS capacitors 201-206 using the drift protectionresistors 321-323. In the illustrated configuration, the DC bias voltageat the anodes of the MOS capacitors 201-206 can be controlled to the DCbias voltage of the RF input RF_(IN) and the RF output RF_(OUT).

Additional details of the variable capacitor cell 450 can be similar tothose described earlier.

Although FIGS. 9A-21B illustrate implementations MOS capacitors usingn-type MOS (NMOS) capacitors, the teachings herein are also applicableto configurations using p type MOS (PMOS) capacitors.

Additionally, although various embodiments of variable capacitor cellsare shown in FIGS. 9A-21B, the teachings herein are also applicable tovariable capacitor cells including a different combination of features.For example, to achieve a desired performance for a particularapplication and/or manufacturing process, a variable capacitor cell caninclude any suitable combination of features of the embodiments of FIGS.9A-21B.

FIG. 22 is a schematic diagram of a cross section of an IC 3002according to one embodiment. The IC 3002 includes a support substrate3010, a buried oxide (BOX) layer 3020 over the support substrate 3010,and a device layer 3030 over the BOX layer 3020. The IC 3002 furtherincludes a substrate contact 3040, which has been provided through theBOX layer 3020 and the device layer 3030 to provide electrical contactto the support substrate 3010.

The illustrated IC 3002 further includes a first MOS capacitor 3110 aand a second MOS capacitor 3110 b. The first MOS capacitor 3110 aincludes source and drain regions 3210 a, 3210 b, respectively, whichcollectively operate as the first MOS capacitor's cathode. The first MOScapacitor 3110 a further includes a first gate region 3230 a, which isdisposed over a first gate oxide region 3220 a and which operates as thefirst MOS capacitor's anode. The second MOS capacitor 3110 b includessource and drain regions 3210 c, 3210 d, respectively, whichcollectively operate as the second MOS capacitor's cathode. The secondMOS capacitor 3110 b further includes a second gate region 3230 b, whichis disposed over a second gate oxide region 3220 b and which operates asthe second MOS capacitor's anode.

In the illustrated configuration, isolation regions have been used tohelp isolate the first and second MOS capacitors 3110 a, 3110 b from oneanother and from other structures of the IC 3002. For example, the firstMOS capacitor 3110 a is positioned between the first and secondisolation regions 3250 a, 3250 b, and the second MOS capacitor 3110 b ispositioned between the second and third isolation regions 3250 b, 3250c. In certain configurations, isolation regions can be used to surrounda perimeter of the first and second MOS capacitors 3110 a, 3110 b whenviewed from above. In one embodiment, the first and second MOScapacitors 3110 a, 3110 b are associated with two different variablecapacitor arrays.

Despite inclusion of the isolation regions 3250 a 3250 c, parasiticcircuit components can result in parasitic coupling between the firstand second MOS capacitors 3110 a, 3110 b. For example, a first parasiticcapacitor CPAR1 can be present between the cathode of the first MOScapacitor 3110 a and the BOX layer 3020 and/or the support substrate3010, and a second parasitic capacitor CPAR2 can be present between thecathode of the second MOS capacitor 3110 b and the BOX layer 3020 and/orthe support substrate 3010. Additionally, the first and second parasiticcapacitors CPAR1, CPAR2 can be electrically connected to one another viaa parasitic resistor RPAR, which can be associated with a resistance ofthe BOX layer 3020 and/or the support substrate 3010.

Although a capacitance of the first and second parasitic capacitorsCPAR1, CPAR2 can be relatively small and a resistance of the parasiticresistor RPAR can be relatively large, parasitic coupling cannevertheless be present between the first and second MOS capacitors 3110a, 3110 b. The parasitic coupling can lead to a degradation of theQ-factor of variable capacitor array that includes the first and secondMOS capacitors 3110 a, 3110 b.

The IC 3002 has been annotated to include the substrate bias circuit3120, which can be used to control a voltage level of the supportsubstrate 3010. For clarity of the figures, the substrate bias circuit3120 has been illustrated schematically as a box. However, the substratebias circuit 3120 can be fabricated on the IC 3002.

In certain configurations, the substrate bias circuit 3120 can be usedto control the voltage level of the support substrate 3010 so as toincrease a resistivity of the parasitic resistor RPAR relative to aconfiguration in which the support substrate 3010 is unbiased orelectrically floating. For example, positive fixed charge in the BOXlayer 3020 can attract electrons to an interface between the BOX layer3020 and the support substrate 3010, which can lead to an inversion oraccumulation layer at the interface. The inversion layer can have aresistance that is much smaller than a resistance of the BOX layer 3020,and thus can serve to increase parasitic coupling between the first andsecond MOS capacitors 3110 a, 3110 b, which can degrade Q-factor.

By biasing the support substrate 3010 using the substrate bias circuit3120, the inversion layer at the interface between the support substrate3010 and the BOX layer 3020 can become depleted. Accordingly, aparasitic interaction between the first and second MOS capacitors 3110a, 3110 b can decrease, and a Q-factor of a variable capacitor arrayincluding the first and second MOS capacitors 3110 a, 3110 b canincrease.

In one embodiment, the substrate bias circuit 3120 is used to controlthe voltage level of the support substrate 3010 to a voltage level inthe range of about 10 V to about 40 V. However, other voltage levels arepossible, including, for example, voltage levels associated with aparticular fabrication process.

Although FIG. 22 illustrates an IC fabricated using an SOI process, theteachings herein are applicable to ICs fabricated using any of a widerange of processing technologies, including, for example, CMOSprocesses.

FIG. 23A is a cross section of a MOS capacitor 3500 according to oneembodiment. The MOS capacitor 3500 includes source and drain regions3510 a, 3510 b, respectively, which collectively operate as the MOScapacitor's cathode. The MOS capacitor 3500 further includes a gateregion 3530, which operates as the MOS capacitor's anode.

As shown in FIG. 23A, the source and drain regions 3510 a, 3510 b aredisposed in the device layer 3030. Additionally, the device layer 3030is disposed over the BOX layer 3020, which in turn is disposed over thesupport substrate 3010. Additionally, the gate oxide region 3520 isdisposed over the device layer 3030, and the gate region 3530 isdisposed over the gate oxide region 3520.

The illustrated MOS capacitor 3500 includes a first halo or pocketimplant 3550 a and a second halo or pocket implant 3550 b. Certainmanufacturing processes include halo implantation to control transistorperformance for relatively small gate lengths, such as gate lengths of50 nm or less. For instance, halo implants can be used to limit anamount of diffusion of source and/or drain regions underneath edges of agate during high temperature processes associated with semiconductorfabrication. Absent inclusion of halo implants, source and drain regionsmay diffuse unduly close to one another. For example, the source anddrain regions may diffuse to provide a relatively short channel lengththat is susceptible to punch through at low drain to source voltage(VDS) voltage levels.

The halo implants can include a doping polarity that is opposite that ofactive regions associated with source and drain regions. For example,when active regions associated with source and drain regions are n type,the halo implants can be p type. Additionally, when active regionsassociated with source and drain regions are p type, the halo implantscan be n type.

FIG. 23B is a cross section of a MOS capacitor 3600 according to anotherembodiment. The MOS capacitor 3600 of FIG. 23B is similar to the MOScapacitor 3500 of FIG. 23A, except that the MOS capacitor 3600 omits thefirst and second halo regions 3550 a, 3550 b of FIG. 23A.

Configuring the MOS capacitor 3600 in this manner can result in arelatively large amount of diffusion of the source and drain regions3510 a, 3510 b. However, in the illustrated configuration, the sourceand drain regions 3510 a, 3510 b are electrically connected to oneanother and operate as a cathode. Thus, the MOS capacitor 3600 canremain operable even when the source and drain regions 3510 a, 3510 bdiffuse relatively close to one another and/or diffuse into one another.

In certain embodiments, a MOS capacitor fabricated without halo orpocket implants can exhibit higher Q-factor and/or smaller capacitancevariation in the presence of RF signals relative to a configuration inwhich the pocket implants are included.

Although FIGS. 23A and 23B illustrate MOS capacitors in context of anSOI process, the teachings herein are applicable to MOS capacitorsfabricated using a wide range of processing technologies, including, forexample, CMOS processes.

Terms such as above, below, over and so on as used herein refer to adevice orientated as shown in the figures and should be construedaccordingly. It should also be appreciated that because regions within asemiconductor device are defined by doping different parts of asemiconductor material with differing impurities or differingconcentrations of impurities, discrete physical boundaries betweendifferent regions may not actually exist in the completed device butinstead regions may transition from one to another. Some boundaries asshown in the accompanying figures are of this type and are illustratedas abrupt structures merely for the assistance of the reader. In theembodiments described above, p-type regions can include a p-typesemiconductor material, such as boron, as a dopant. Further, n-typeregions can include an n-type semiconductor material, such asphosphorous, as a dopant. A skilled artisan will appreciate variousconcentrations of dopants in regions described above.

CONCLUSION

Unless the context clearly requires otherwise, throughout thedescription and the claims, the words “comprise,” “comprising,” and thelike are to be construed in an inclusive sense, as opposed to anexclusive or exhaustive sense; that is to say, in the sense of“including, but not limited to.” The word “coupled”, as generally usedherein, refers to two or more elements that may be either directlyconnected, or connected by way of one or more intermediate elements.Likewise, the word “connected”, as generally used herein, refers to twoor more elements that may be either directly connected, or connected byway of one or more intermediate elements. Additionally, the words“herein,” “above,” “below,” and words of similar import, when used inthis application, shall refer to this application as a whole and not toany particular portions of this application. Where the context permits,words in the above Detailed Description using the singular or pluralnumber may also include the plural or singular number respectively. Theword “or” in reference to a list of two or more items, that word coversall of the following interpretations of the word: any of the items inthe list, all of the items in the list, and any combination of the itemsin the list.

Moreover, conditional language used herein, such as, among others,“can,” “could,” “might,” “can,” “e.g.,” “for example,” “such as” and thelike, unless specifically stated otherwise, or otherwise understoodwithin the context as used, is generally intended to convey that certainembodiments include, while other embodiments do not include, certainfeatures, elements and/or states. Thus, such conditional language is notgenerally intended to imply that features, elements and/or states are inany way required for one or more embodiments or that one or moreembodiments necessarily include logic for deciding, with or withoutauthor input or prompting, whether these features, elements and/orstates are included or are to be performed in any particular embodiment.

The above detailed description of embodiments of the invention is notintended to be exhaustive or to limit the invention to the precise formdisclosed above. While specific embodiments of, and examples for, theinvention are described above for illustrative purposes, variousequivalent modifications are possible within the scope of the invention,as those skilled in the relevant art will recognize. For example, whileprocesses or blocks are presented in a given order, alternativeembodiments may perform routines having steps, or employ systems havingblocks, in a different order, and some processes or blocks may bedeleted, moved, added, subdivided, combined, and/or modified. Each ofthese processes or blocks may be implemented in a variety of differentways. Also, while processes or blocks are at times shown as beingperformed in series, these processes or blocks may instead be performedin parallel, or may be performed at different times.

The teachings of the invention provided herein can be applied to othersystems, not only the system described above. The elements and acts ofthe various embodiments described above can be combined to providefurther embodiments.

While certain embodiments of the inventions have been described, theseembodiments have been presented by way of example only, and are notintended to limit the scope of the disclosure. Indeed, the novel methodsand systems described herein may be embodied in a variety of otherforms; furthermore, various omissions, substitutions and changes in theform of the methods and systems described herein may be made withoutdeparting from the spirit of the disclosure. The accompanying claims andtheir equivalents are intended to cover such forms or modifications aswould fall within the scope and spirit of the disclosure.

What is claimed is:
 1. A capacitor structure comprising: a substrate; asource/drain region formed in said substrate to form an active areahaving an active area width, said source/drain region having areflection symmetry; and a plurality of gates formed above saidsubstrate, each of said plurality of gates having a gate width, saidgate width configured to be less than said active area width, and saidplurality of gates formed to have reflection symmetry.
 2. The capacitorstructure of claim 1, wherein said substrate is an silicon-on-insulatorsubstrate.
 3. The capacitor structure of claim 1, wherein said pluralityof source/drain regions, and said plurality of gates are interconnectedto form one or more pairs of capacitors connected in an anti-seriesconfiguration.
 4. The capacitor structure of claim 1, wherein saidplurality of source/drain regions, and said plurality of gates areinterconnected to form a variable capacitor cell of a variable capacitorarray.
 5. The capacitor structure of claim 4, wherein the variablecapacitor cell is part of an integrated circuit.
 6. The capacitorstructure of claim 1, wherein said plurality of source/drain regions,and said plurality of gates are interconnected to form a plurality ofvariable capacitor cells of a variable capacitor array.
 7. A method toform a plurality of capacitors comprising: forming a source/drain regionin a substrate to form an active area having an active area width, saidsource/drain region having reflection symmetry; and forming a pluralityof gates above said substrate, each of said plurality of gates having agate width, said gate width configured to be less than said active areawidth, and said plurality of gates formed to have reflection symmetry.8. The method of claim 7, wherein said substrate is ansilicon-on-insulator substrate.
 9. The method of claim 7, furthercomprising forming connections between said plurality of source/drainregions, and said plurality of gates to form one or more pairs ofcapacitors connected in an anti-series configuration.
 10. The method ofclaim 7, further comprising forming connections between said pluralityof source/drain regions, and said plurality of gates to form a variablecapacitor cell of a variable capacitor array.
 11. The capacitorstructure of claim 7, further comprising forming connections betweensaid plurality of source/drain regions, and said plurality of gates toform a plurality of variable capacitor cells of a variable capacitorarray.
 12. An integrated circuit comprising: a substrate; a source/drainregion formed in said substrate to form an active area having an activearea width, said source/drain region having an active area width; and aplurality of gates formed above said substrate, each of said pluralityof gates having a gate width, said gate width configured to be less thansaid active area width, and said plurality of gates formed to havereflection symmetry.
 13. The integrated circuit of claim 12, whereinsaid substrate is a silicon-on-insulator substrate.
 14. The integratedcircuit of claim 12, wherein said plurality of source/drain regions, andsaid plurality of gates are interconnected to form one or more pairs ofcapacitors connected in an anti-series configuration.
 15. The integratedcircuit of claim 12, wherein said plurality of source/drain regions, andsaid plurality of gates are interconnected to form a variable capacitorcell of a variable capacitor array.
 16. The integrated circuit of claim12, wherein said plurality of source/drain regions, and said pluralityof gates are interconnected to form a plurality of variable capacitorcells of a variable capacitor array.
 17. The integrated circuit of claim12, further comprising a bias voltage generator configured to generate abias voltage for each one of said plurality of variable capacitor cellsof said variable capacitor array.
 18. The integrated circuit of claim17, further comprising an interface configured to receive a controlsignal for said bias voltage generator used to adjust a value of saidbias voltage for each one of said plurality of variable capacitor cellsof said variable capacitor array.
 19. The integrated circuit of claim18, wherein said interface is a Mobile Industry Processor Interfaceradio front end interface.
 20. The integrated circuit of claim 12,wherein said plurality of source/drain regions, and said plurality ofgates are interconnected to form a plurality of variable capacitorarrays.